Spectral stitching method to increase instantaneous bandwidth in vector signal analyzers

ABSTRACT

Various embodiments are described of devices and associated methods for processing a signal using a plurality of vector signal analyzers (VSAs). An input signal may be split and provided to a plurality of VSAs, each of which may process a respective frequency band of the signal, where the respective frequency bands have regions of overlap. Each VSA may adjust the gain and phase of its respective signal such that continuity of phase and magnitude is preserved through the regions of overlap. The correction of gain and phase may be accomplished by a complex multiply with a complex calibration constant. A complex calibration constant may be determined for each VSA by comparing the gain and phase of one or more calibration tones generated with each region of overlap, as measured by each of the VSAs.

PRIORITY INFORMATION

This application is a continuation of U.S. patent application Ser. No.15/072,909, entitled “Spectral Stitching Method to IncreaseInstantaneous Bandwidth in Vector Signal Analyzers”, whose inventors areStephen L. Dark, Daniel J. Baker, and Johnathan R. W. Ammerman, filedMar. 17, 2016; which is a continuation of U.S. patent application Ser.No. 14/515,144, entitled “Spectral Stitching Method to IncreaseInstantaneous Bandwidth in Vector Signal Analyzers”, whose inventors areStephen L. Dark, Daniel J. Baker, and Johnathan R. W. Ammerman, filedOct. 15, 2014, now U.S. Pat. No. 9,326,174 issued on Apr. 26, 2016; allof which are incorporated herein by reference in its entirety as thoughfully and completely set forth herein.

FIELD OF THE INVENTION

The present invention relates to the field of signal processing, andmore particularly to systems and methods for increasing instantaneousbandwidth in a vector signal analyzer or a vector signal generator.

DESCRIPTION OF THE RELATED ART

Instantaneous bandwidth is an important banner specification for allradio frequency (RF) vector signal analyzers (VSAs) and RF vector signalgenerators (VSGs). The desire of the industry is to increase thebandwidth as much as possible without sacrificing dynamic range. In manycases, the limiting factor in achieving the largest possible bandwidthis the sample rate of the analog-to-digital converters (ADCs) anddigital-to-analog converters (DACs). While ADC and DAC vendors arealways working to increase the converter rates, there still exists adesire in many applications for bandwidths that exceed the capabilitiesof state-of-the-art ADCs and DACs. There are several different methodswithin the industry for achieving larger bandwidths, each with their owndisadvantages.

The prior methods for creating larger instantaneous bandwidths that donot intentionally sacrifice resolution fall into two categories: (1)Time-Interleaving (time) and (2) Quadrature Mixing (phase). Both ofthese methods are industry standards for increasing bandwidth.

Time-Interleaving uses converters that have larger bandwidths than theirsample rates allow. For ADCs, this is made possible by placing fastersample-and-hold circuits on the front-end than the ADC backend iscapable of digitizing. Then, by taking N ADCs and staggering them intime by the ADC sample period divided by N, the samples can beinterleaved together to create an effective larger sample rate. Thismethod introduces errors resulting from inaccuracies in staggering thetime alignment and from differences in the magnitude and phase betweenthe various ADCs. Therefore, several online and offline DSP correctionalgorithms have been created to combat these effects. In general, it isdifficult to achieve more than 8 bits of dynamic range without DSPcorrection. With offline DSP correction, this can be improved to betterthan 12 bits of dynamic range but can be very sensitive to temperature.Online methods typically have to assume something about the inputsignal, and other negative effects occur when those assumptions arebroken.

Quadrature mixing uses a quadrature down-converter or up-converter tomix an RF signal into or from two signals, an in-phase signal and aquadrature-phase signal. In the case of a down-converter, the RF signalis mixed with a sinusoid to create the in-phase signal and the same RFsignal is mixed with a sinusoid that is 90 degrees out of phase with thein-phase sinusoid to create the quadrature phase signal. Finally, eachof these two analog signals is digitized with ADCs. These two signalsare typically represented as a single complex signal, where the in-phasesignal represents the real part and the quadrature-phase part of thesignal represents the imaginary part. As a result, the positivefrequency bandwidth is independent of the negative frequency bandwidth.Thus the net effect is a doubling of the bandwidth. Using this method,the full resolution of the data converters is preserved. However, thismethod typically creates a DC leakage spur and an image spur. Inaddition, the method only scales to two converts.

Thus, there exists a need for mechanisms capable of achieving the goalof larger instantaneous or modulation bandwidths from smaller bandwidthswithout the scalability and image rejection issues of the quadraturemixing technique and without the inaccuracies present for timeinterleaving methods.

SUMMARY

Methods and systems are disclosed for processing a signal using aplurality of vector signal analyzers (VSAs). In a presented method, eachof the plurality of VSAs may be provided with a respective componentsignal comprising a copy of a respective frequency band of an inputsignal. The combination of the respective frequency bands may comprisean aggregate frequency band having an aggregate center frequency. Eachrespective frequency band may have a respective region of overlap withat least one other respective frequency band. Each respective frequencyband may also have a respective center frequency with a respectivefrequency offset from the aggregate center frequency. Each of the VSAsmay be phase-locked and time-synchronized with respect to the otherVSAs.

Each of the VSAs may process the provided respective component signal.The processing may comprise digitizing, interpolating,frequency-shifting, filtering, and adjusting the gain and phase of therespective component signal. The digitizing may comprise shifting therespective center frequency of the respective frequency band tobaseband, and sampling at least a portion of the respective componentsignal corresponding to the respective measurement band. Thefrequency-shifting may comprise shifting the respective component signalsuch that the respective center frequency is offset from baseband by therespective frequency offset. The filtering the respective componentsignals may be configured to cause a sum of the component signals tohave a unity frequency response within each region of overlap. Thefiltering may be performed using a digital half-band filter. Theadjusting gain and phase of the respective component signals may beconfigured to cause the sum of the component signals to have acontinuous frequency response over the aggregate frequency band.

The method may further comprise summing the respective component signalsto obtain a composite signal.

A system is presented for processing a signal. The apparatus maycomprise a signal splitter configured to receive an analog signal, andoutput a plurality of copies of the analog signal. The system mayfurther comprise a plurality of output ports, each of the communicationports configured to provide to a respective vector signal analyzer (VSA)a respective copy of the modulated signal. The system may furthercomprise a plurality of input ports, each of the input ports configuredto receive from the respective VSA a respective digital signal. Eachrespective digital signal may comprise a digitized version of arespective frequency band of the analog signal. Each respectivefrequency band may have a region of overlap with at least one otherfrequency band received by another of the communication ports. Eachrespective frequency band may also have a respective center frequencyhaving a respective frequency offset from an aggregate center frequencyof an aggregate frequency band. The aggregate frequency band maycomprise the combination of the frequency bands of the received digitalsignals. The respective VSA may be phase-locked and time-synchronizedwith respect to the VSAs of the other input ports.

The system may further comprise a plurality of parallel signalprocessing pathways. Each of the parallel signal processing pathways maybe configured to receive a respective digital signal from one of theinput ports, and digitize, interpolate, frequency-shift, filter, andadjust the gain and phase of the respective digital signal according tothe method described above.

The system may further comprise a summing unit configured to sum theoutputs of the plurality of parallel signal processing pathways toobtain a composite signal.

A method is provided for calibrating a signal processing systemincluding at least a first VSA and a second VSA. The method may compriseproviding a first component signal to the first VSA and providing asecond component signal to the second VSA. The first component signalmay comprise a first frequency band within an aggregate frequency bandof an input signal, and the second component signal may comprise asecond frequency band within the aggregate frequency band of the inputsignal. The aggregate frequency band may have an aggregate centerfrequency. The first frequency band may have a first center frequency ata first frequency offset from the aggregate center frequency, and thesecond frequency band may have a second center frequency at a secondfrequency offset from the aggregate center frequency. The secondfrequency band may have a region of overlap with the first frequencyband, the region of overlap containing a calibration tone. The secondVSA may be phase-locked and time-synchronized with respect to the firstVSA.

The method may further comprise digitizing, interpolating,frequency-shifting, and filtering each of the first and second componentsignals, according to the method described above. The method may furthercomprise computing a complex calibration constant based on a magnituderatio and a phase difference. The magnitude ratio may be determined by amagnitude of the calibration tone measured by the first VSA and amagnitude of the calibration tone measured by the second VSA. The phasedifference may be determined by a phase of the calibration tone measuredby the first VSA and a phase of the calibration tone measured by thesecond VSA.

The method may further comprise storing the complex calibration constantin memory. The complex calibration constant may be useable to correctphase and gain mismatch between the first VSA and the second VSA.

BRIEF DESCRIPTION OF THE DRAWINGS

A better understanding of the present inventions can be obtained whenthe following detailed description is considered in conjunction with thefollowing drawings:

FIG. 1 is a block diagram illustrating an embodiment of a system forperforming spectral stitching in a receive path;

FIG. 2 is a block diagram illustrating an embodiment of parallel vectorsignal analyzers (VSAs);

FIGS. 3a and 3b illustrate an exemplary embodiment of respectivefrequency bands within an aggregate frequency band with and withoutcalibration tones;

FIG. 4 illustrates the signal response of a half-band filter;

FIG. 5 is a block diagram illustrating another embodiment of a systemfor performing spectral stitching in a receive path;

FIGS. 6a and 6b illustrate measurements of a calibration tone, asperformed by two VSAs before and after phase and magnitude adjustment,represented in the time domain;

FIG. 7 is a block diagram illustrating an embodiment of a system forperforming spectral stitching in a transmit path; and

FIG. 8 is a block diagram illustrating an embodiment of a digital signalprocessing block for use in the system of FIG. 7.

While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof are shown by way ofexample in the drawings and are herein described in detail. It should beunderstood, however, that the drawings and detailed description theretoare not intended to limit the invention to the particular formdisclosed, but on the contrary, the intention is to cover allmodifications, equivalents and alternatives falling within the spiritand scope of the present invention as defined by the appended claims.Note that the various section headings in the following DetailedDescription are for organizational purposes only and are not meant to beused to limit the claims.

DETAILED DESCRIPTION Incorporation by Reference

The following references are incorporated by reference as if fully andcompletely disclosed herein:

U.S. Patent Application No. 2013/0343490, filed Jun. 20, 2012, entitled“Synchronizing Receivers in a Signal Acquisition System”, invented byWertz et al.;

U.S. Pat. No. 7,624,294, issued on Nov. 24, 2009, entitled“Synchronizing Measurement Devices Using Trigger Signals”, invented byCraig M. Conway; and

U.S. Pat. No. 7,315,791, issued on Jan. 1, 2008, entitled “ApplicationProgramming Interface for Synchronizing Multiple InstrumentationDevices”, invented by Kosta Ilic et al.

Terminology

The following is a glossary of terms used in the present application:

Memory Medium—Any of various types of memory devices or storage devices.The term “memory medium” is intended to include an installation medium,e.g., a CD-ROM, floppy disks 105, or tape device; a computer systemmemory or random access memory such as DRAM, DDR RAM, SRAM, EDO RAM,Rambus RAM, etc.; a non-volatile memory such as a Flash, magnetic media,e.g., a hard drive, or optical storage; registers, or other similartypes of memory elements, etc. The memory medium may comprise othertypes of memory as well or combinations thereof. In addition, the memorymedium may be located in a first computer in which the programs areexecuted, or may be located in a second different computer whichconnects to the first computer over a network, such as the Internet. Inthe latter instance, the second computer may provide programinstructions to the first computer for execution. The term “memorymedium” may include two or more memory mediums which may reside indifferent locations, e.g., in different computers that are connectedover a network.

Programmable Hardware Element—includes various hardware devicescomprising multiple programmable function blocks connected via aprogrammable interconnect. Examples include FPGAs (Field ProgrammableGate Arrays), PLDs (Programmable Logic Devices), FPOAs (FieldProgrammable Object Arrays), and CPLDs (Complex PLDs). The programmablefunction blocks may range from fine grained (combinatorial logic or lookup tables) to coarse grained (arithmetic logic units or processorcores). A programmable hardware element may also be referred to as“reconfigurable logic”.

Computer System—any of various types of computing or processing systems,including a personal computer system (PC), mainframe computer system,workstation, network appliance, Internet appliance, personal digitalassistant (PDA), television system, grid computing system, or otherdevice or combinations of devices. In general, the term “computersystem” can be broadly defined to encompass any device (or combinationof devices) having at least one processor that executes instructionsfrom a memory medium.

Local Oscillator (LO)—a circuit configured to generate a periodic signalat a specified frequency and amplitude. The periodic signal may be apure sinusoid, and its frequency and/or amplitude may be programmable.The periodic signal may or may not be phase or frequency locked toanother periodic signal.

Overview

Time interleaving uses time as the mechanism to increase the bandwidthwhile quadrature mixing uses phase as its mechanism. The present“spectral stitching” approach uses frequency as its mechanism to achievelarger instantaneous bandwidths. The spectral stitching approach may beapplied to both signal receivers, such as RF (radio frequency)receivers, for example, and signal generators, such as RF generators,for example, as discussed below. As used herein, the term “RF” isintended to include the full spectrum of communication frequencies, andincludes at least radio and microwave frequencies.

Embodiments of the present invention may be realized in any of variousforms. For example, in some embodiments, the present invention may berealized as a computer-implemented method, a computer-readable memorymedium, or a computer system. In other embodiments, the presentinvention may be realized using one or more custom-designed hardwaredevices such as ASICs. In other embodiments, the present invention maybe realized using one or more programmable hardware elements such asFPGAs.

In some embodiments, a computer-readable memory medium may be configuredso that it stores program instructions and/or data, where the programinstructions, if executed by a computer system, cause the computersystem to perform a method, e.g., any of a method embodiments describedherein, or, any combination of the method embodiments described herein,or, any subset of any of the method embodiments described herein, or,any combination of such subsets.

In some embodiments, a computer system may be configured to include aprocessor (or a set of processors) and a memory medium, where the memorymedium stores program instructions, where the processor is configured toread and execute the program instructions from the memory medium, wherethe program instructions are executable to implement any of the variousmethod embodiments described herein (or, any combination of the methodembodiments described herein, or, any subset of any of the methodembodiments described herein, or, any combination of such subsets). Thecomputer system may be realized in any of various forms. For example,the computer system may be a personal computer (in any of its variousrealizations), a workstation, a computer on a card, anapplication-specific computer in a box, a server computer, a clientcomputer, a hand-held device, a tablet computer, a wearable computer,etc.

Receive Path

In a receive path, spectral stitching may be performed by using aplurality N of vector signal analyzers (VSAs) to digitize an analoginput receive (RX) signal, such as an RF signal, where each VSA handlesa respective frequency band of the signal. Together the respectivefrequency bands comprise an aggregate frequency band of interest. Theoutputs of the N VSAs may therefore be recombined to form a compositesignal having a bandwidth on the order of N times the bandwidth of eachindividual VSA, thus covering the entire aggregate frequency band. Theaggregate frequency band may be a region of interest within the input RXsignal.

FIG. 1 illustrates a block diagram of an embodiment of a system 100 forperforming spectral stitching in a signal path receiving an inputRXsignal. As shown in FIG. 1, the system 100 includes three VSAs 108 a-c.Other embodiments may include another number N of VSAs. It should beappreciated that the terms “VSA” and “vector signal analyzer,” as usedherein, may encompass any type of signal analyzer, digitizer, receiver,or other device capable of converting, or configured to convert, ananalog input signal to a digital output signal.

One or more calibration tones may be added to the inputRX signal to aidin calibrating the plurality of VSAs, as discussed below. In someembodiments, the one or more calibration tones may be generated bycalibration tone generator 104, and added to the input signal by a powercombiner 102. In other embodiments, the one or more calibration tonesmay be added to the input signal using a simple two-port mux, or usingany other signal combining technique known in the art. In someembodiments, the system 100 may include multiple calibration tonegenerators, which may require the power combiner 102 to have more thantwo inputs. The one or more calibration tones may be generated and addedto the input signal while the system 100 is operating in a calibrationmode, as discussed below. Thus, in normal operation, the power combiner102 may not add the one or more calibration tones to the input signal.

A copy of at least a portion of the inputRX signal may be provided toeach of the VSAs 108 a-c. This may be accomplished by splitting theinput signal, such as by using a power splitter 106, having N outputs.

The output of each VSA 108 a-c may be provided to a summing unit 110.The output of the summing unit 110 is a composite signal representingthe sum of the output of each of the VSAs 108 a-c.

As mentioned previously, each of the VSAs 108 a-c may handle a differentfrequency band of the inputRX signal. To ensure continuity across thefull aggregate frequency band, the different frequency bands handled bythe respective VSAs 108 a-c should overlap. Thus, in order for the sumof the outputs of the VSAs 108 a-c to accurately represent a digitizedversion of the aggregate frequency band of the inputRX signal, theoutputs of the VSAs 108 a-c may be further processed to providecontinuity through the regions of overlap. Each of the VSAs 108 a-c maytherefore comprise signal processing capabilities beyond thosetraditionally included in a VSA. In some embodiments, each of the VSAs108 a-c may not comprise a stand-alone VSA. For example, each of theVSAs 108 a-c may be implemented as a signal processing path on aprogrammable hardware element, multiprocessor system, etc.

FIG. 2 illustrates a block diagram providing further detail of anembodiment of the VSAs 108 a-c. Specifically, blocks 202 a-210 arepresent details of an embodiment of VSA 108 a, blocks 202 b-210 brepresent details of an embodiment of VSA 108 b, and blocks 202 c-210 crepresent details of an embodiment of VSA 108 c. Summing unit 110 isalso included in FIG. 2 for context.

At each of the digitize blocks 202 a-c, a component signal comprising arespective frequency band of the inputRX signal may be digitized. Eachof the digitize blocks 202 a-c may include functionality included in atraditional VSA.

In some embodiments, each of the digitize blocks 202 a-c may receive acopy of the entire inputRX signal, e.g. from the signal splitter 106. Inother embodiments, each of the digitize blocks 202 a-c may receive onlya respective portion of the inputRX signal. Because each of the VSAs 108a-c may handle a different frequency band of the inputRX signal, each ofthe digitize blocks 202 a-c may digitize a respective component signalcomprising a respective frequency band, each respective frequency bandidentified by a respective center frequency.

FIG. 3a illustrates an exemplary embodiment of frequency bands asreceived by the digitize blocks 202 a-c, represented in the frequencydomain. In this example, frequency band 302 represents the frequencyband of the digitize block 202 a (i.e. VSA 108 a), frequency band 304represents the frequency band of the digitize block 202 b (i.e. VSA 108b), and frequency band 306 represents the frequency band of the digitizeblock 202 c (i.e. VSA 108 c). The region covered by frequency bands304-306 together represents the aggregate frequency band identified byan aggregate center frequency. Each respective center frequency isoffset from the aggregate center frequency by a respective frequencyoffset. In some circumstances a respective center frequency may beoffset from the aggregate center frequency by 0 Hz, as in the example offrequency band 304.

The frequency bands may overlap to avoid gaps within the aggregatefrequency band. For example, the region 308 represents a region ofoverlap between frequency band 302 and frequency band 304 (i.e. betweenthe respective component signals of VSAs 108 a and 108 b), and theregion 310 represents the region of overlap between frequency band 304and frequency band 306 (i.e. between the respective component signals ofVSAs 108 b and 108 c).

The digitizing performed by each of the digitize blocks 202 a-c maycomprise performing I/Q demodulation on the respective component signalto produce a pair of analog I (in-phase) and Q (quadrature) signals.

The digitizing performed by each of the digitize blocks 202 a-c may alsocomprise frequency-shifting the respective component signal (or the I/Qsignal pair) such that the respective center frequency is shifted tobaseband. Each of the digitize blocks 202 a-c may then filter outportions of the shifted signal that are outside the respective frequencyband, e.g. by using a low-pass filter. Alternatively, each of thedigitize blocks 202 a-c may frequency-shift the received inputRX signalto a position other than baseband (or forego frequency-shifting), andfilter the shifted signal using a band-pass filter.

In one embodiment, each of the digitize blocks 202 a-c may comprise arespective local oscillator (LO) operating at the respective centerfrequency. The respective LO may be used, for example, infrequency-shifting the respective center frequency to baseband. The LOsof the digitize blocks 202 a-c may have a fixed phase differencerelative to each other. For example, the LOs may be locked to a commonreference, such that the relative phases between the devices will remainfixed.

The digitizing performed by each of the digitize blocks 202 a-c mayfurther comprise complex sampling the filtered signal, as known in theart. The VSAs 108 a-c may be time-synchronized, such that the respectivesignals may be sampled simultaneously in each of the digitize blocks 202a-c. Alternatively, the respective signals may be sampled at aconsistent offset of time. In this case, the consistent offset may bemeasured and corrected. Each of the digitize blocks 202 a-c may output acomplex (I/Q) signal.

At filter blocks 204 a-c, the respective component signals may befiltered. Because the respective frequency bands of the respectivecomponent signals overlap in frequency, as shown in FIG. 3, the overlapregions should be filtered to prevent power spikes, or other artificialincreases in magnitude, in the overlap regions when the respectivesignals are summed by summing unit 110. In other words, the respectivecomponent signals should be filtered such that their sum appearscontinuous. Specifically, the respective component signals may befiltered such that the sum of overlapping signals provides a unityresponse at all points within the aggregate frequency band. Moregenerally, this continuous-sum filtering may be configured in any mannersuch that the summed signals approximate the result that would beachieved if the entire aggregate frequency band had been digitized by asingle VSA having sufficient bandwidth to digitize the entire aggregatefrequency band.

Various filter shapes may be used to accomplish this. For example, FIG.4 illustrates the response of a half-band filter, where the solid trace402 represents a filter response for a first VSA and the dotted trace404 represents a filter response for a second VSA with an overlappingfrequency band. In FIG. 4, the crossover point is located at 30 MHzwhere there is a 10 MHz crossover region. While a half-band filterinherently has the needed spectral characteristics to filter overlappingfrequency bands to sum together to produce unity gain, it forces thecrossover point to occur at the sampling frequency divided by four,fs/4. In other embodiments, this crossover may be moved further out infrequency using other filter methods, e.g., to increase theeffectiveness of the natural instantaneous bandwidth of each device.

The filtering illustrated as filter blocks 204 a-c may happen at any ofvarious points in the VSA. For example, the filtering may occur afterthe interpolate blocks 206 a-c. Alternatively, some embodiments mayperform the filtering of the filter blocks 204 a-c within the digitizeblocks 202 a-c, e.g., concurrently with the low-pass filtering of thedigitize blocks 202 a-c. In this case, the filtering may be performed byan analog filter prior to complex sampling of the filtered signal.

At interpolate blocks 206 a-c, each respective component signal may beinterpolated. The interpolation factor should be set such that eachrespective interpolate block 206 interpolates the respective componentsignal to at least the effective I/Q rate required for the “stitched”data's bandwidth. For example, in one embodiment, the effective I/Q ratemay be required to be at least the Nyquist rate of the respectivecomponent signal. In another embodiment, a higher rate (e.g., 1.25 timesthe Nyquist rate) may be selected.

At frequency shift blocks 208 a-c, each respective component signal maybe shifted into the proper location in frequency relative to the otherdevices. As a result, each device will frequency shift its interpolatedspectrum to a different location. Specifically, each respectivecomponent signal may be shifted such that its respective centerfrequency is offset from baseband by the respective frequency offset bywhich it was originally offset from the aggregate center frequency.Thus, the entire aggregate frequency band is frequency shifted to centerat baseband.

For example, in the case that there are three VSAs each using half-bandfilters with an I/Q rate of 120 MHz, then the cross-over points will belocated at positive and negative 30 MHz. This means that the threerespective center frequencies may be shifted to [−60 MHz, 0 Hz, 60 MHz].Thus, the respective frequency bands should be defined such that therespective frequency offsets are [−60 MHz, 0 Hz, 60 MHz] relative to theaggregate center frequency.

If the digitize blocks 202 a-c previously shifted the respective centerfrequencies to baseband, then VSA 108 a may, in this example, frequencyshift its spectrum to the left by 60 MHz, VSA 108 b may shift by 0 Hz,and VSA 108 c may frequency shift its spectrum to the right by 60 MHz.In other words, each respective center frequency may be shifted by itsrespective frequency offset. In embodiments in which the respectivefrequencies were shifted by the digitize blocks 202 a-c to a frequencyother than baseband, then the respective center frequencies may beshifted by some value other than the respective frequency offsets.

At the gain and phase correction blocks 210 a-c, the magnitude and phaseof each respective component signal may be adjusted to make the spectrumcontinuous through the regions of overlap. This gain and phasecorrection may comprise a complex multiply of each of one or more of therespective component signals with a respective calibration constant.Determining a calibration constant for each of the gain and phasecorrection blocks 210 a-c is discussed below.

Where the VSAs 108 a-c are not time-synchronized, but have a constantdelay relative to each other, the gain and phase correction blocks 210a-c may be further configured to measure and correct the delay.

FIG. 5 illustrates a block diagram of an embodiment of a second system500 for performing spectral stitching in a signal path receiving aninputRX signal. As shown in FIG. 5, the system 500 includes three VSAs508 a-c. Other embodiments may include another number N of VSAs.

In FIG. 5, the power combiner 102, the calibration tone generator 104,and the power splitter 106 may operate as described with regard toFIG. 1. The VSAs 508 a-c may be standard VSAs as known in the art,without the additional signal processing capabilities of VSAs 108 a-c.Instead, the additional signal processing functions may be performed bya separate digital signal processing block 510. For example, the digitalsignal processing block 510 may perform the functions of filtering,interpolating, frequency shifting, and gain and phase correction, asdiscussed with regard to FIG. 2, blocks 204-210, for each of the VSAs508 a-c. For example the digital signal processing block 510 maycomprise a separate signal processing path for processing the output ofeach of the VSAs 508 a-c. The digital signal processing block 510 mayfurther comprise a summing function, which may function in a mannersimilar to the summing unit 110.

The system 500 presents an advantage over the system 100, in that thesystem 500 may allow a user to utilize standard, off-the-shelf VSAs. Forexample, system 500 may be realized in the form of a signal processingchassis comprising the digital signal processing block 510, andoptionally further comprising one or more of the power combiner 102, thecalibration tone generator 104, and the power splitter 106. The signalprocessing chassis may further comprise slots to accept a plurality ofVSAs, which may be standard, off-the-shelf VSAs. The signal processingchassis may be configured to operate with a variable number of VSAs,according to the preferences of the user. Further, the VSAs may differin bandwidth, resolution, or other characteristics, according to thepreferences of the user.

Determining VSA Calibration Constants

In order to adjust the magnitudes and phases of the respective signalsin the receive path to provide continuity through the regions ofoverlap, the relative magnitudes and phases between the respectivesignals without adjustment may be determined. This may be performed byinjecting a calibration tone at each crossover point, or region ofoverlap, of the respective frequency bands. The calibration tone maythen be measured and compared by the VSAs. Differences and/or ratiosbetween the measurements by different VSAs of the magnitude and phase ofthe calibration tone may be used to determine calibration constants foreach of the VSAs.

For example, to determine calibration constants for a system such as thesystem 100 shown in FIGS. 1-2, or the system 500 shown in FIG. 5, thesystem may be set to a calibration mode. The calibration mode will bediscussed herein with respect to the system 100 shown in FIGS. 1-2.However, the same principles may be applied to other embodiments, suchas the system 500.

As shown in FIG. 1, the power combiner 102 may be used to combine theinput signal with one or more calibration tones from the calibrationtone generator 104. A calibration tone may comprise a real tone at asingle known frequency falling within a region of overlap. A calibrationtone may also have a known magnitude. In some embodiments, a calibrationtone may be injected at a single region of overlap at a time, with eachregion of overlap being treated sequentially. In other embodiments,calibration tones may be injected at multiple, or all, regions ofoverlap simultaneously. The combined signal may then be used as theinput to the splitter 106 where each of the splitter's outputs may bethe input to one of N VSAs, such as VSAs 108 a-c.

At the N VSAs, each of the N−1 regions of overlap may comprise arespective calibration tone, as shown in FIG. 3b (either simultaneouslyor sequentially). As illustrated, calibration tone 312 may be generatedwithin region of overlap 308 (i.e. within the respective signals of VSAs108 a and 108 b), and calibration tone 314 may be generated withinregion of overlap 310 (i.e. within the respective signals of VSAs 108 band 108 c).

At blocks 202-208, each respective signal may be digitized, filtered,interpolated, and frequency shifted, as discussed above with regard toFIG. 2. However, the gain and phase correction block 210 may operatedifferently in the calibration mode. Specifically, the gain and phasecorrection block 210 may measure the respective calibration tone withineach region of overlap. For example, the gain and phase correction block210 a (of VSA 108 a) may measure the calibration tone 312, since itfalls within the respective frequency band 302, which is processed bythe VSA 108 a. The gain and phase correction block 210 b (of VSA 108 b)may also measure the calibration tone 312, since it also falls withinthe respective frequency band 304, which is processed by the VSA 108 b.

FIG. 6a illustrates exemplary results of measurements of the calibrationtone 312, as performed by the VSAs 108 a and 108 b, represented in thetime domain. As illustrated, the in-phase (I) and quadrature (Q)components of the calibration tone 312 as measured by the VSA 108 a havea phase and magnitude that are different from the phase and magnitude ofthe I and Q components of the calibration tone 312 as measured by theVSA 108 b. This may result from normal differences in the hardware,temperature, etc. of the VSAs 108 a and 108 b. In this condition, thesum of the respective signals output by the VSAs 108 a and 108 b willnot be continuous through the region of overlap, because of the mismatchin phase and magnitude.

Using the measurements of the calibration tone within each region ofoverlap, respective calibration constants may be determined, for use inrealigning the respective signals through each region of overlap. Forexample, for each respective VSA, a complex calibration constant may bedetermined that, when complex multiplied by a calibration tone measuredby that respective VSA, will result in an output calibration tone havinga phase and magnitude matching an output calibration tone of an adjacentVSA. FIG. 6b illustrates output calibration tones of VSAs 108 a and 108b. The output calibration tones shown in FIG. 6b represent the signalsshown in FIG. 6a after being multiplied by the determined calibrationconstants.

In one embodiment, the magnitudes of the calibration tones generated bythe calibration tone generator 104 may be known. The calibrationconstants may therefore be determined such that the output calibrationtones have a magnitude matching the generated calibration tones. Inanother embodiment, the calibration constants may merely be determinedsuch that the output calibration tones produced by adjacent VSAs havethe same magnitude.

For example, calibration tone 312 may be measured by both VSA 108 a andVSA 108 b. Calibration tone 314 may be measured by both VSA 108 b andVSA 108 c. The output signal of VSA 108 a will include an outputcalibration tone corresponding to calibration tone 312. The outputsignal of VSA 108 b will include output calibration tones correspondingto calibration tone 312 and calibration tone 314. The output signal ofVSA 108 c will include an output calibration tone corresponding tocalibration tone 314.

A first calibration constant may optionally be determined for VSA 108 asuch that complex multiplication of the first calibration constant withthe output signal of VSA 108 a results in the output calibration tonecorresponding to calibration tone 312 having a magnitude matching theknown magnitude of calibration tone 312.

A second calibration constant may be determined for VSA 108 b such thatcomplex multiplication of the second calibration constant with theoutput signal of VSA 108 b results in the output calibration tonecorresponding to calibration tone 312 having a phase matching the phaseof the output calibration tone of VSA 108 a. The second calibrationconstant may further be determined such that the output calibration tonecorresponding to calibration tone 312 has a magnitude matching the knownmagnitude of calibration tone 312 and/or the magnitude of the outputcalibration tone of VSA 108 a corresponding to calibration tone 312. Inone embodiment, a calibration constant determined to cause the outputcalibration tone of 108 b to match the phase and magnitude of the outputcalibration tone of VSA 108 a may be determined by performing a complexdivision of the calibration tone 312 as measured by VSA 108 a by thecalibration tone 312 as measured by VSA 108 b. Because the complexmultiplication of the second calibration constant is performed with theentire output signal of VSA 108 b, the output calibration tone of VSA108 b corresponding to calibration tone 314 is also adjusted.

A third calibration constant may be determined for VSA 108 c such thatcomplex multiplication of the third calibration constant with the outputsignal of VSA 108 c results in the output calibration tone correspondingto calibration tone 314 having a phase matching the phase of theadjusted output calibration tone of VSA 108 b corresponding tocalibration tone 314. The third calibration constant may further bedetermined such that the output calibration tone corresponding tocalibration tone 314 has a magnitude matching the known magnitude ofcalibration tone 314, and/or the magnitude of the output calibrationtone of VSA 108 b corresponding to calibration tone 314.

The calibration constants for each of the VSAs should be determined, orre-determined, each time the phases of the VSAs change relative to eachother. For example, the relative phases of the VSAs may change if therelative phases of the LOs of the VSAs change. This may occur, e.g., ifthe LO frequencies of one or more VSAs change and the one or more VSAsare relocked. In some VSAs, the phase of the LO can be made to bedeterministic. If this is the case, then a calibration constant may bedetermined once and stored for each frequency of the VSA. The storedcalibration constant may then be recalled at some future time withoutthe need for recalibration.

Transmit Path

In a transmit path, spectral stitching may be performed by using aplurality N of vector signal generators (VSGs) to generate an outputanalog transmit (TX) signal, such as an RF signal, where each VSGhandles a respective frequency band of the signal. Together therespective frequency bands comprise an aggregate frequency band ofinterest. The outputs of the N VSAs may therefore be combined to form acomposite signal having a bandwidth on the order of N times thebandwidth of each individual VSG, thus covering the entire aggregatefrequency band.

FIG. 7 illustrates a block diagram of an embodiment of a system 700 forperforming spectral stitching in a signal path generating an outputanalog signal. As shown in FIG. 7, the system 700 includes three VSGs704 a-c. Other embodiments may include another number N of VSGs. Itshould be appreciated that the terms “VSG” and “vector signalgenerator,” as used herein, may encompass any type of signal generator,transmitter, or other device capable of converting, or configured toconvert, a digital input signal to an analog output signal.

A digital input signal, illustrated in FIG. 7 as the “TX Signal”, may beprovided to the system. The input TX Signal may comprise a complexdigital signal having a bandwidth that is larger than the bandwidth ofeach respective VSG. Therefore, each of the VSGs 704 a-c may be providedwith a respective component signal comprising at least a portion of theinput TX Signal. Specifically, each component signal may comprise arespective frequency band of the input TX Signal, and each of the VSGs704 a-c may process the respective frequency band. Each respectivefrequency band may have a respective center frequency having arespective frequency offset from an aggregate center frequency of theaggregate frequency band.

To ensure continuity across the full aggregate frequency band, therespective frequency bands should overlap. Thus, in order for theoutputs of the VSGs 704 a-c to be recombined to accurately represent ananalog version of the aggregate frequency band of the digital inputsignal, the component signals may be further processed to providecontinuity through the regions of overlap. Each of the VSGs 704 a-c maytherefore comprise signal processing capabilities beyond thosetraditionally included in a VSG. Alternatively, such further processingmay be performed by signal processing circuitry outside of the VSGs 704a-c, such as by the digital signal processing (DSP) block 702, which isdescribed more fully below. Such an embodiment would allow a user toimplement the present invention using standard off-the-shelf VSGs. Inother embodiments, each of the VSGs 704 a-c may not comprise astand-alone VSG. For example, each of the VSGs 704 a-c may beimplemented as a signal processing path on a programmable hardwareelement, multiprocessor system, etc.

Once the further processing has been performed, e.g. by the DSP block702 or by each of the VSGs 704 a-c, each of the VSGs 704 a-c may convertthe respective component signal to an analog signal. Each of the VSGs704 a-c may also up-convert the respective component signal such thatthe aggregate center frequency is located at a desired carrierfrequency, and each respective center frequency is offset from thedesired carrier frequency by the respective frequency offset.

The combiner 706 may comprise a power combiner with a plurality ofinputs, or may comprise any other hardware for combining analog signals,as known in the art. The combiner 706 may receive as inputs the outputsof the VSGs 704 a-c, and may output a composite signal comprising acombination of its inputs.

A power splitter 708 may provide copies of the composite signal as anoutput of the system, i.e. as “TX Out” shown in FIG. 7, and also to acalibration receiver 710. The calibration receiver 710 may be used toreceive one or more calibration tones for use in determining calibrationconstants for one or more of the VSGs 704 a-c, as discussed below. Thecalibration receiver 710 may be phase locked to the VSGs 704 a-c.

FIG. 8 is a block diagram illustrating the DSP block 702 in greaterdetail. The DSP block 702 may comprise a plurality of parallelprocessing paths, each of which may process one of the respectivecomponent signals. In some embodiments, the DSP block 702 may beseparate from the VSGs 704 a-c, as shown in FIG. 7. In otherembodiments, a respective one of the parallel processing paths of theDSP block 702 may be included in each of the VSGs 704 a-c. For example,blocks 802 a, 804 a, 806 a, and 808 a may be included in VSG 704 a;blocks 802 b, 804 b, 806 b, and 808 b may be included in VSG 704 b; andblocks 802 c, 804 c, 806 c, and 808 c may be included in VSG 704 c.

In some embodiments, each of the frequency shift blocks 802 a-c mayreceive a copy of the entire digital input signal. In other embodiments,each of the frequency shift blocks 802 a-c may receive only a respectiveportion of the digital input signal comprising a respective frequencyband.

FIG. 3a illustrates an exemplary embodiment of frequency bands asreceived by the frequency shift blocks 802 a-c, represented in thefrequency domain. In this example, frequency band 302 represents thefrequency band of the frequency shift block 802 a (i.e. VSG 704 a),frequency band 304 represents the frequency band of the frequency shiftblock 802 b (i.e. VSG 704 b), and frequency band 306 represents thefrequency band of the frequency shift block 802 c (i.e. VSG 704 c). Theregion covered by frequency bands 304-306 together represents theaggregate frequency band identified by an aggregate center frequency.Each respective center frequency is offset from the aggregate centerfrequency by a respective frequency offset. In some circumstances arespective center frequency may be 0 Hz, as in the example of frequencyband 304.

The frequency bands may overlap to avoid gaps within the aggregatefrequency band. For example, the region 308 represents a region ofoverlap between frequency band 302 and frequency band 304 (i.e. betweenthe respective component signals of VSGs 704 a and 704 b), and theregion 310 represents the region of overlap between frequency band 304and frequency band 306 (i.e. between the respective component signals ofVSGs 704 b and 704 c).

At the frequency shift blocks 802 a-c, each of the respective componentsignals may be shifted to baseband. Where the aggregate frequency bandis initially at baseband, this means that each respective componentsignal is frequency-shifted by the negative of the respective frequencyoffset. For example, if the respective frequency offset of therespective component signal being processed by the frequency shift block802 c is 60 MHz, then the frequency shift block 802 c would frequencyshift the respective component signal by −60 MHz.

At the decimate blocks 804 a-c, each respective component signal may bedecimated to a rate that is less than or equal to the maximum samplerate of the corresponding VSG. This decimation may comprise merelydropping samples. Alternatively, this decimation may comprisealias-protected decimation, utilizing an alias protection filter.

The filter blocks 806 a-c are similar to the filter blocks 204 a-c ofFIG. 2. Because the respective frequency bands of the respectivecomponent signals overlap in frequency, as shown in FIG. 3, the overlapregions should be filtered to prevent power spikes in the overlapregions when the respective signals are combined by the combiner 706. Inother words, the respective component signals should be filtered suchthat their sum appears continuous. Specifically, the respectivecomponent signals may be filtered such that the sum of overlappingsignals provides a unity response at all points within the aggregatefrequency band.

Various filter shapes may be used to accomplish this. For example, FIG.4 illustrates the response of a half-band filter, where the solid trace402 represents a filter response for a first VSG and the dotted trace404 represents a filter response for a second VSG with an overlappingfrequency band. In FIG. 4, the crossover point is located at 30 MHzwhere there is a 10 MHz crossover region. While a half-band filterinherently has the needed spectral characteristics to filter overlappingfrequency bands to sum together to produce unity gain, it forces thecrossover point to occur at the sampling frequency divided by four,fs/4. In other embodiments, this crossover may be moved further out infrequency using other filter methods, e.g., to increase theeffectiveness of the natural instantaneous bandwidth of each device.

At the gain and phase correction blocks 808 a-c, the magnitude and phaseof each respective component signal may be adjusted to make the spectrumcontinuous through the regions of overlap. This gain and phasecorrection may comprise a complex multiply of each of one or more of therespective component signals with a respective calibration constant.Determining a calibration constant for each of the gain and phasecorrection blocks 808 a-c is discussed below.

Determining VSG Calibration Constants

In order to adjust the magnitudes and phases of the respective signalsin the transmit path to provide continuity through the regions ofoverlap, the relative magnitudes and phases between the respectivesignals without adjustment may be determined using one or morecalibration tones. This may be performed in multiple ways.

For example, to determine calibration constants for a system such as thesystem 700 shown in FIGS. 7-8, the system 700 may be set to acalibration mode, which may operate according to one of the followingapproaches.

In a first approach, a calibration tone may be added to the input TXSignal within a region of overlap of the respective frequency band ofthe first VSG and the respective frequency band of a second, adjacentVSG. The calibration tone may be generated by a digital calibration tonegenerator (not shown), which may be comprised within the DSP block 702,or may be a separate component. The calibration tone may be added to theinput TX Signal using a switch, a multiplexer, or by using any othermethod known in the art. Preferably, the calibration tone may begenerated at the center of the region of overlap. For example, thecalibration tone generator may generate a calibration tone 312 withinthe region 308, as shown in FIG. 3b . The respective outputs of the VSG704 a and the VSG 704 b will thus each include a representation of thecalibration tone. With the output of the VSG 704 b turned off ordisabled, the magnitude and phase of the representation of thecalibration tone 312 present in the output of the VSG 704 a may bemeasured by the calibration receiver 710. The output of the first VSGmay then be turned off or disabled, and the output of the second VSG maybe enabled. For example, the VSG 704 b may be enabled. Therepresentation of the calibration tone 312 present in the output of theVSG 704 b may then be measured by the calibration receiver 710. Wherethe calibration receiver 710 is phase-locked with the VSGs 704 a and 704b throughout the time when the two measurements are made, a relativephase difference of the VSGs 704 a and 704 b may be determined bycomparing the phases of the two representations of the calibration tone312, as measured by the calibration receiver 710. A relative magnitudedifference of the VSGs 704 a and 704 b may also be determined bycomparing the magnitudes of the two representations of the calibrationtone 312. A calibration constant may then be determined for one or moreof the VSGs 704 a and 704 b, based on the determined relative phasedifference and the determined relative magnitude difference. This methodmay be repeated for each of the regions of overlap.

In a second approach, an iterative method may be used to determine therelative magnitude and phases. In this method, both the VSG 704 a andthe VSG 704 b may each simultaneously generate an output comprising arepresentation of the calibration tone 312. Then, the magnitude andphase of the VSG 704 b may be iteratively adjusted, seeking to force therespective representations of the calibration tone 312 present in therespective outputs of VSG1 and VSG2 to deconstructively interfere. Inother words, the magnitude and phase of the representation of thecalibration tone 312 present in the output of the VSG 704 b may beiteratively adjusted until the total output power at the frequency ofthe representation of the calibration tone 312 is minimized. Thecalibration constant may then be determined by negating the VSG 704 bresult to rotate it by 180 degrees. This procedure may then be repeatedfor each overlap region. While this second approach would take longerthan the direct measurement of the first approach, this second approachdoes not require the calibration receiver 710 to be phase locked to theVSGs. Moreover, since the calibration receiver 710 is only makingunlocked power measurements, the calibration receiver 710 may bereplaced by a power meter, thereby simplifying the hardware requirementsof the calibration circuitry.

The calibration constants for each of the VSGs should be determined, orre-determined, each time the phases of the VSGs change relative to eachother. For example, the relative phases of the VSGs may change if therelative phases of the LOs of the VSGs change. This may occur, e.g., ifthe LO frequencies of one or more VSGs change and the one or more VSGsare relocked. In some VSGs, the phase of the LO can be made to bedeterministic. If this is the case, then a calibration constant may bedetermined once and stored for each frequency of the VSG. The storedcalibration constant may then be recalled at some future time withoutthe need for recalibration.

Although the embodiments above have been described in considerabledetail, numerous variations and modifications will become apparent tothose skilled in the art once the above disclosure is fully appreciated.It is intended that the following claims be interpreted to embrace allsuch variations and modifications.

We claim:
 1. An apparatus for processing a signal, the apparatuscomprising: a first signal processing pathway configured to: receive afirst component signal comprising a first frequency band of an inputsignal; and digitize the first component signal; a second signalprocessing pathway, phase-locked and time-synchronized with respect tothe first signal processing pathway, the second signal processingpathway configured to: receive a second component signal comprising asecond frequency band of the input signal, the second frequency bandhaving a region of overlap with the first frequency band; and digitizethe second component signal; wherein, when the second signal processingpathway is in a calibration mode, the second signal processing pathwayis further configured to compute a complex calibration constant, whereinthe complex calibration constant is computed based on a phase differencebetween a first digitized version of a calibration tone output of by thefirst signal processing pathway and a second digitized version of thecalibration tone output by the second signal processing pathway, whereinthe calibration tone is included within the region of overlap betweenthe first frequency band and the second frequency band; and a memoryconfigured to store the complex calibration constant.
 2. The apparatusof claim 1, wherein, when the second signal processing pathway is not inthe calibration mode, the second signal processing pathway is furtherconfigured to: correct, via application of the stored complexcalibration constant, phase mismatch between a digitized output of thefirst signal processing pathway and a digitized output of the secondsignal processing pathway, using the stored calibration constant.
 3. Theapparatus of claim 1, wherein the complex calibration constant iscomputed further based on a ratio of a magnitude of the first digitizedversion of the calibration tone and a magnitude of the second digitizedversion of the calibration tone.
 4. The apparatus of claim 1, whereinthe first signal processing pathway is further configured to: filter thefirst component signal; and wherein the second signal processing pathwayis further configured to: filter the second component signal; whereinsaid filtering of the first and second component signals is configuredto cause a sum of the first and second component signals to have a unityfrequency response within the region of overlap.
 5. The apparatus ofclaim 4, wherein said filtering is performed by one or more half-bandfilters.
 6. The apparatus of claim 1, wherein said digitizing the firstcomponent signal comprises shifting the first component signal such thatthe first frequency band is at baseband, and said digitizing the secondcomponent signal comprises shifting the second component signal suchthat the second frequency band is at baseband; wherein the second signalprocessing pathway is further configured to: frequency-shift thedigitized second component signal such that a frequency offset betweenthe digitized first component signal and the digitized second componentsignal is the same as a frequency offset between the first frequencyband of the input signal and the second frequency band of the inputsignal.
 7. The apparatus of claim 6, wherein the first signal processingpathway is further configured to: interpolate the first componentsignal, wherein said interpolating is performed after said digitizingthe first component signal and before said frequency-shifting thedigitized first component signal; wherein the second signal processingpathway is further configured to: interpolate the second componentsignal, wherein said interpolating is performed after said digitizingthe second component signal and before said frequency-shifting thedigitized second component signal.
 8. An apparatus for processing asignal, the apparatus comprising: a first signal processing pathway thatis phase-locked and time-synchronized with respect to a second signalprocessing pathway, the first signal processing pathway configured to:receive a first component signal comprising a first frequency band of aninput signal, the first frequency band having a region of overlap with asecond frequency band of the input signal; digitize the first componentsignal; compute a complex calibration constant, wherein the complexcalibration constant is computed based on a phase difference between afirst digitized version of a calibration tone output by the first signalprocessing pathway and a second digitized version of the calibrationtone output by the second signal processing pathway when an input of thesecond signal processing pathway comprises the second frequency band,wherein the calibration tone is included within the region of overlapbetween the first frequency band and the second frequency band; and amemory configured to store the complex calibration constant.
 9. Theapparatus of claim 8, wherein the first signal processing pathway isfurther configured to: correct, via application of the stored complexcalibration constant, phase mismatch between a digitized output of thefirst signal processing pathway and a digitized output of the secondsignal processing pathway, using the stored calibration constant. 10.The apparatus of claim 8, wherein the complex calibration constant iscomputed further based on a ratio of a magnitude of the first digitizedversion of the calibration tone and a magnitude of the second digitizedversion of the calibration tone.
 11. The apparatus of claim 8, whereinthe first signal processing pathway is further configured to: filter thefirst component signal, wherein said filtering the first componentsignal is configured to cause a sum of the output of the first signalprocessing pathway and the output of the second signal processingpathway to have a unity frequency response within the region of overlap.12. The apparatus of claim 11, wherein said filtering is performed byone or more half-band filters.
 13. The apparatus of claim 8, whereinsaid digitizing the first component signal comprises shifting the firstcomponent signal such that the first frequency band is at baseband;wherein the first signal processing pathway is further configured to:frequency-shift the digitized first component signal such that afrequency offset between the output of the first signal processingpathway and the output of the second signal processing pathway is thesame as a frequency offset between the first frequency band of the inputsignal and the second frequency band of the input signal.
 14. Theapparatus of claim 13, wherein the first signal processing pathway isfurther configured to: interpolate the first component signal, whereinsaid interpolating is performed after said digitizing the firstcomponent signal and before said frequency-shifting the digitized firstcomponent signal.
 15. A method for processing a received signal, themethod comprising: digitizing each of a first component signalcomprising a first frequency band of the received signal and a secondcomponent signal comprising a second frequency band of the receivedsignal, the first frequency band and the second frequency band having aregion of overlap, wherein a first digitized version of a calibrationtone is included in the digitized first component, and wherein a secondversion of the calibration tone is included in the digitized secondcomponent; computing a complex calibration constant, wherein the complexcalibration constant is computed based on a phase difference between thefirst digitized version of the calibration tone and the second digitizedversion of the calibration tone, wherein the calibration tone isincluded within the region of overlap between the first frequency bandand the second frequency band; and storing the complex calibrationconstant.
 16. The method of claim 15, further comprising: correcting,via application of the stored complex calibration constant, phasemismatch between the digitized first component signal and the digitizedsecond component signal, using the stored calibration constant.
 17. Themethod of claim 15, wherein the complex calibration constant is computedfurther based on a ratio of a magnitude of the first digitized versionof the calibration tone and a magnitude of the second digitized versionof the calibration tone.
 18. The method of claim 15, further comprising:filtering each of the first component signal and the second componentsignal, wherein said filtering of the first and second component signalsis configured to cause a sum of the first and second component signalsto have a unity frequency response within the region of overlap.
 19. Themethod of claim 18, wherein said filtering is performed by one or morehalf-band filters.
 20. The method of claim 15, wherein said digitizingeach of the first component signal and the second component signalcomprises shifting the first component signal such that the firstfrequency band is at baseband, and shifting the second component signalsuch that the second frequency band is at baseband; the method furthercomprising: frequency-shifting at least one of the digitized firstcomponent signal and the digitized second component signal, such that afrequency offset between the digitized first component signal and thedigitized second component signal is the same as a frequency offsetbetween the first frequency band of the first component signal and thesecond frequency band of the second component signal.